The return loss of a connector system can be adversely affected when the impedance (of two conductors forming a transmission line) through the connector is not. This connector impedance can be affected by the compensation network that is applied within a jack, and in certain instances, the compensation network can increase the return loss.
Crosstalk in a plug-jack connector system can manifest itself as NEXT (Near End Crosstalk) and FEXT (Far End Crosstalk). A schematic cross-sectional view of connecting hardware components, and a communications signal path through them, is shown in FIGS. 1A and 1B. In particular, FIG. 1A illustrates a plug 100 connected to a jack 105 both of which have communication cables 110,115 attached thereto, respectively. The jack 105 includes a printed circuit board (PCB) 120 with insulation displacement contacts (IDCs) 125 and plug interface contacts (PICs) 130. The communication signal passing through the connecting hardware is illustrated in FIG. 1B as the dotted line 135.
Associated with the plug 100 there exists a known amount of offending crosstalk (set by an ANSI/TIA (American National Standards Institute/Telecommunications Industry Association) standard) between any two wire-pairs. This offending crosstalk may be canceled or reduced by a compensating signal within the jack 105. In order to cancel or reduce the offending crosstalk, a compensating signal that is approximately 180° out of phase with the offending plug-crosstalk may need to be injected. Because of the propagation delay between the plug's 100 offending crosstalk signal and the compensating signal that is injected within the jack 105, the two signals cannot totally cancel each other in the frequency range of interest. There will be a remaining uncompensated signal that will limit the performance of the system with respect to the NEXT performance specification.
FIG. 2 shows a generalized example of a prior-art connecting hardware system where crosstalk 140 occurs and a single compensation stage 145 is implemented. The coupling (that creates the offending crosstalk and compensation signals) in connectors arises from capacitive and/or mutual inductive coupling. The coupling magnitude is dependent on the physical construction and dimensions of the signal conductors, the material properties, and the magnitude of the signal. This coupling is also proportional to frequency (approximately 20 dB/decade). An equivalent representation of the coupling in the connecting hardware of FIG. 2 is shown in FIG. 3. The 90° (offending crosstalk signal 50) and the −90° (compensation signal 55) coupling are shown with reference to the source signal traveling on the sour wire-pair 60.
The source signal energy propagates from coupling location A to coupling location B (propagation time=T/2), couples to the sink wire-pair 65 (forming the compensation signal), and then propagates back to coupling location A (having another propagation time of T/2) on the sink wire-pair with a resultant time delay T. FIG. 4 illustrates a lumped approximation of the magnitude and polarity for the offending crosstalk vector A and the compensation vector B on a time axis.
The round trip time delay T is due to distance between the coupling location A and the coupling location B, and the signal's velocity. While this time delay is fixed, the compensating signal's phase difference of 180° (at very low frequencies) increases at higher frequencies. The magnitude of each coupling will typically increase with frequency as well (e.g., at a 20 dB per decade slope). A complex vectorial summation of the two signals A and B creates the remaining uncompensated signal which results in the NEXT signal. FIG. 5 is a vector diagram in polar form of the offending crosstalk vector A, the compensating vector B, and the resultant signal (i.e., NEXT vector) for a typical connecting hardware with a typical distance and materials between offending plug-crosstalk and compensating coupling positions. In order to be able to present small coupling signals simultaneously with large coupling signals on the same diagram, the magnitude of the couplings is presented in logarithmic (dB) scale relative to the source signal. The five dots represent the vectors' magnitudes at the following frequencies: 1, 10, 100, 250, and 500 MHz.
By choosing the vectors A and B of equal magnitudes with reverse polarity (i.e., approximately 180° out of phase), the vectors A and B's combined crosstalk will be approximately zero, or at least relatively negligible, only at low frequencies. This is because at such low frequencies the phase difference between the A and B vectors is close to 180°. However, at higher frequencies the phase difference grows, resulting in a bigger combined NEXT magnitude. For this reason, the physical distance between the offending crosstalk in the plug and the compensation can be important. For a fixed signal velocity, the closer the plug coupling crosstalk position (A) is to the compensation coupling position (B), the higher the possible bandwidth the connector design will have (due to the smaller phase difference).
By using this basic single-stage method (with conventional materials and dimensions) the crosstalk can be maintained at an acceptable level up to approximately 100 MHz resulting in a connector that will comply with Category 5e (ANSI/TIA-568-C.2) requirements for NEXT. A typical NEXT signal (the resultant signal) as a function of frequency for an existing single-stage-compensation system is illustrated in FIG. 6.
To achieve a superior NEXT performance level at higher frequencies, multiple-compensation-stage methods have been introduced by the industry. An example of such a multiple-compensation-stage technique is shown in FIG. 7, which illustrates an additional compensation stage 150 that was added after (with respect to distance or time) the first compensating stage 145. In this case, the magnitude of the first compensating coupling 145 needs to be adjusted to offset the additional compensation stage 150.
An equivalent diagram of the coupling in the connecting hardware of FIG. 7 is shown in FIG. 8, and a lumped approximation of the signal magnitude and polarity for the offending crosstalk vector A, first stage compensation vector B, and additional compensation vector C on a time axis is shown in FIG. 9. The magnitudes and phases of these vectors relative to each other is illustrated in FIG. 10 on a polar axis plot. The magnitudes of the couplings are presented in logarithmic (dB) scale relative to the source signal. The five dots again represent the vectors' magnitudes at the following frequencies: 1, 10, 100, 250, and 500 MHz.
With vector A's location as a reference, with increasing frequency, vector B's phase shift will grow clockwise towards vector A, and vector C's phase shift will grow clockwise more swiftly (due to its location further away from A) in opposition to vector A. Selecting |B| equal to |A+C| at a given a frequency, requires that |B|<|A+C| below that given frequency. To demonstrate the occurrence more clearly, FIG. 11 shows a typical combined crosstalk performance through frequency for a two-stage-compensation jack. The approach is largely relying on the phase of A+C being equal or very close to the coupling B's phase over the entire frequency spectrum of interest for the connector.
The magnitude of |A+C| is greater than the magnitude of |B| in the low end of the frequency bandwidth. At a certain (predetermined) frequency, the magnitude of |A+C| will be equal to the magnitude of |B| (creating a minimum as shown in FIG. 11). At the higher end of the frequency bandwidth of the connector, the magnitude of |A+C| will be smaller than magnitude 1131, resulting in an increasing resultant magnitude. This prior art design technique improves the frequency bandwidth of the connector by reducing the phase delay's effect on the first stage compensation vector.
Another example of multiple-stage compensation is illustrated in FIG. 12. The order of compensation couplings are as follows: a first compensation stage 145 providing compensating coupling; a second compensating stage 150 providing crosstalk coupling; and then a third compensation stage 155 providing compensating coupling. The sum of the numeric magnitude of the crosstalk couplings (offending and compensating crosstalk) and compensating coupling needs to be close to each other. An equivalent diagram of the three-stage compensating coupling in a connecting hardware is shown in FIG. 13, and a lumped approximation of the magnitude and polarity of the occurring vectors is shown in FIG. 14 along a time a time axis. In particular, FIG. 14 shows vector A representing coupling from the offending crosstalk, vector B representing the coupling from the first compensation stage, vector C representing the coupling from the second compensation stage, and vector D representing the coupling from the third compensation stage.
In order for a three-stage compensation technique to work, the flowing conditions should be valid: (i) the magnitude of the offending crosstalk coupling A is close to the magnitude of compensating coupling D; (ii) the magnitude of the compensating crosstalk coupling C is close to the magnitude of compensating coupling B; (iii) the combined magnitude of the couplings B and C are greater than combined magnitude of couplings A and D; and (iv) the numeric summation of coupling A and coupling C is approximately equal to numeric summation of coupling B and coupling D.
FIG. 15 illustrates the vector magnitudes and phases relative to each other in a polar axis for a three-stage compensation technique. The magnitudes of the couplings are presented in logarithmic (dB) scale relative to source signal. The five dots present the vectors' positions at 1, 10, 100, 250, and 500 MHz. The summation of the vectors of this three-stage compensation technique over a frequency range is shown further in FIG. 16. At one frequency the phase shift of vectors (A+C) and (B+D) will obtain exact opposite polarity (180° out of phase) which will drive the summation of all vectors (i.e., |A|+|B|+|C|+|D|) to take a dip in magnitude.
The various multi-stage compensation methods described above generally require additional coupling stages with more overall coupling. This can make connectors which employ such compensation techniques more sensitive to tolerances in manufacturing processes. Additionally, due to the high coupling magnitude of the compensation vectors a wire-pair's impedance will likely be affected resulting in an impedance mismatch with the cable and a poor return loss. Improved compensation techniques for use in network connectors are desired.